1. Introduction
As an emerging clean energy source, high-capacity power batteries are being widely used in electric vehicles, communication systems, aerospace, and other fields. The formation process is a critical step in battery production, directly affecting the quality and cost of the battery. However, the repeated charge and discharge cycles during formation consume a large amount of electrical energy, especially for large-capacity EV battery packs. The charging and discharging modules in conventional formation equipment operate under low-voltage and high-current conditions, which makes it difficult to improve the overall efficiency. This paper focuses on the design of both the discharge and charge modules for high-capacity EV battery pack formation equipment. A novel transformer series-parallel topology is proposed for the discharge module, while a current-doubler synchronous rectification topology is adopted for the charge module. System parameters are designed, and prototype experiments are carried out to verify the feasibility and efficiency of the proposed solutions.
2. Design of the Discharge Module
2.1 Topology Analysis and Proposed Structure
Typical isolated DC-DC converter topologies include forward, flyback, push-pull, half-bridge, and full-bridge converters. Their main features are compared in Table 1.
| Topology | Advantages | Disadvantages | Power Range |
|---|---|---|---|
| Forward | Simple circuit, low cost, reliable drive | Unidirectional transformer excitation, low utilization | Hundreds of watts to several kW |
| Flyback | Very simple circuit, low cost | Hard to reach high power, low transformer utilization | Hundreds of watts to tens of watts |
| Push-pull | Bidirectional transformer excitation, simple drive | Potential flux imbalance | Hundreds of watts to several kW |
| Half-bridge | Bidirectional excitation, fewer switches | DC bias issue, complex isolated drive | Hundreds of watts to several kW |
| Full-bridge | Bidirectional excitation, high power capability | Complex structure, high cost | Several kW to hundreds of kW |
For the discharge module, the input is low voltage and high current, while the output is high voltage and low current. The push-pull topology is selected as the basic cell because of its simple drive and high transformer utilization. To further increase the input current capability and output voltage level, three identical push-pull converters are connected in parallel at the input and in series at the output as shown in the network representation below.
The composite network can be described by its transmission matrix. For two sub-networks connected in parallel at input and series at output, the overall \(ABCD\) matrix is the sum of individual matrices:
$$
\begin{bmatrix}
A & B \\
C & D
\end{bmatrix}
=
\begin{bmatrix}
A_1 & B_1 \\
C_1 & D_1
\end{bmatrix}
+
\begin{bmatrix}
A_2 & B_2 \\
C_2 & D_2
\end{bmatrix}
$$
If the two sub-converters have identical external characteristics, each converter carries half of the total input current and produces half of the total output voltage. This automatically balances the power sharing, eliminating the need for dedicated current-sharing control. Moreover, the current rating of each switching device is halved, and the transformer turns ratio as well as its size can be reduced.
2.2 Main Circuit Design
The main circuit of the discharge module consists of three push-pull converters with transformer series-parallel connection. The key design parameters are listed in Table 2.
| Parameter | Value |
|---|---|
| Battery discharge voltage (input) | 1.0 – 3.6 V (typical EV battery pack cell) |
| DC bus voltage (output) | 300 V |
| Discharge current | 10 – 100 A |
| Current ripple | < 5% |
| Efficiency at 100 A | > 90% |
2.3 High-Frequency Transformer Design
The transformer is the core component of the isolated converter. Key considerations include core material, winding arrangement, and loss minimization. Ferrite material with high resistivity is selected for high-frequency operation. The core geometry coefficient \(K_g\) method is used to determine the transformer size.
The apparent power \(P_t\) is:
$$
P_t = P_o \left( \frac{1}{\eta} + 1 \right)
$$
The electrical condition coefficient \(K_e\) is given by:
$$
K_e = 0.145 \cdot K_f^2 \cdot f^2 \cdot B_{ac}^2 \cdot 10^{-4}
$$
The core geometry coefficient is:
$$
K_g = \frac{P_t}{2 \cdot K_e \cdot \alpha}
$$
where \(\alpha\) is the regulation factor. The primary turns \(N_p\) are calculated using Faraday’s law:
$$
N_p = \frac{V_{in(min)} \cdot D_{max} \cdot 10^8}{K_f \cdot f \cdot B_{ac} \cdot A_c}
$$
The secondary turns \(N_s\) are:
$$
N_s = N_p \cdot \frac{V_s}{V_p}
$$
The actual transformer used in the prototype was wound with copper foil for the secondary and litz wire for the primary, using a sandwich winding technique to reduce leakage inductance and skin effect losses.
2.4 Output Rectifier and Filter Design
The output rectifier uses a full-bridge diode configuration. Since the output voltage is high (300 V) and the current is relatively low, fast-recovery diodes are appropriate. The diode stress is equal to the total output voltage divided by the number of transformer series modules. The output filter inductor \(L_f\) must keep the current continuous above 10% of full load:
$$
L_f = \frac{V_{out}}{2 \cdot f_s \cdot I_{out(min)}} \cdot \left( 1 – \frac{2 \cdot D_{min}}{1} \right)
$$
The output capacitor is chosen based on the acceptable ripple voltage \(\Delta V_{ripple}\) and the equivalent series resistance (ESR):
$$
C_f = \frac{I_{ripple(p-p)} \cdot ESR}{\Delta V_{ripple}}
$$
Two 470 µF / 250 V electrolytic capacitors in series are used to achieve the required voltage rating and acceptable ripple.
2.5 Control Circuit Design
The control circuit uses a push-pull PWM controller with built-in dead-time control and soft-start. The switching frequency is set by an external resistor and capacitor. The dead-time is adjusted to prevent shoot-through. The error amplifier is operated as a voltage follower, and the output drives two MOSFETs through a high-current push-pull stage. The control circuit also includes input undervoltage lockout and pulse-by-pulse current limiting.
3. Design of the Charge Module
3.1 Topology Selection
The charge module converts the DC bus voltage (300 V) down to a low voltage (1.0 – 3.6 V) with high current (10 – 100 A) to charge the EV battery pack. A half-bridge converter is selected as the primary stage because it offers balanced transformer excitation and lower switch voltage stress. For the secondary side, the current-doubler synchronous rectification topology is chosen. Its main advantages are:
- Single secondary winding without center tap
- Two output inductors share the output current, reducing copper loss
- Interleaved inductor currents cancel ripple, reducing output filter size
- Synchronous rectifiers significantly reduce conduction loss compared to Schottky diodes
The working principle of the current-doubler synchronous rectifier is illustrated by the following key waveforms over one switching cycle. During the on-time of the upper switch, one inductor stores energy while the other freewheels through its synchronous rectifier; during the off-time, both inductors freewheel through the two synchronous rectifiers. The output current ripple cancellation factor \(K_{ripple}\) depends on the duty cycle \(D\):
$$
K_{ripple} = \frac{1 – 2D}{1 – D}
$$
When \(D\) is close to 0.5, the ripple cancellation is maximized.
3.2 System Parameter Design
The specifications of the charge module are given in Table 3.
| Parameter | Value |
|---|---|
| Input DC bus voltage | 300 V |
| Output battery voltage | 1.0 – 3.6 V |
| Charging current | 10 – 100 A |
| Current ripple | < 5% |
| Efficiency at 100 A | > 90% |
3.2.1 Transformer Design for the Charge Module
The transformer is designed using the same core geometry method as the discharge module. The design conditions are listed in Table 4.
| Design condition | Value |
|---|---|
| Topology | Half-bridge |
| Minimum input voltage | 250 V |
| Maximum input voltage | 350 V |
| Output current | 0 – 100 A |
| Output power | 360 W (maximum) |
| Switching frequency | 100 kHz |
| Efficiency | 90% |
| Regulation | 0.5% |
| Flux density swing | 0.1 T |
The primary and secondary turns are calculated similarly:
$$
N_p = \frac{V_{in(min)}}{2 \cdot f_s \cdot B_{ac} \cdot A_c} \cdot D_{max}
$$
For the present design, \(N_p = 20\) turns and \(N_s = 1\) turn. The secondary winding uses a single layer of copper foil to handle the high current and minimize conduction loss.
3.2.2 Output Filter Inductors
Two identical inductors are used in the current-doubler configuration. The inductance value is determined by the allowable ripple current \(\Delta I_L\). With an output current ripple of ±20% at full load:
$$
L = \frac{V_{out}}{\Delta I_L \cdot f_s} \cdot (1 – D)
$$
Choosing \(f_s = 100\) kHz, \(V_{out} = 3.6\) V, and \(\Delta I_L = 20\) A, the calculated inductance is about \(1.8\ \mu\)H. An actual inductance of \(2\ \mu\)H is selected.
3.2.3 Blocking Capacitor and Input Capacitors
To prevent flux imbalance in the half-bridge transformer, a series blocking capacitor is used. The capacitance value is based on the allowed voltage dip \(\Delta V_c\):
$$
C = \frac{I_{p(max)} \cdot T_{on}}{\Delta V_c}
$$
where \(I_{p(max)}\) is the maximum primary current and \(T_{on}\) is the maximum on-time. A 1 µF / 250 V film capacitor is selected.
The input divider capacitors are chosen to maintain a stable midpoint voltage. Two electrolytic capacitors of 470 µF / 200 V are used.
3.2.4 Power Switch and Synchronous Rectifier Selection
The primary switching devices are MOSFETs with a voltage rating of 500 V and a current rating of 20 A, providing adequate margin for the 300 V bus. The synchronous rectifiers are low-voltage MOSFETs with a very low \(R_{DS(on)}\) to minimize conduction loss. Their ratings are 40 V and 120 A. Table 5 summarizes the selected devices.
| Device | Part | Voltage rating | Current rating | \(R_{DS(on)}\) |
|---|---|---|---|---|
| Primary MOSFET | IRFP450A | 500 V | 20 A | 0.4 Ω |
| Synchronous rectifier | IRL3803 | 40 V | 120 A | 0.006 Ω |
3.3 Control and Driving Circuit
The charging control adopts a two-stage profile: constant current (CC) with a voltage limit, followed by constant voltage (CV) with a current limit. The control system diagram is shown in the block diagram above. Two PI regulators are used in parallel, and the lower output is selected as the control reference. During the initial charging phase, the voltage PI saturates and the current PI regulates the charging current. When the battery voltage reaches the limit, the voltage PI takes over and the current gradually decreases, completing the CC-CV transition automatically.
The driving circuit uses a PWM controller with complementary outputs. Gate drive transformers provide the isolation and level shifting for the primary switches. The synchronous rectifiers are also driven by the same PWM outputs through logic gates and drivers, ensuring proper turn-on and turn-off timing. A dead time is inserted between the complementary signals to prevent shoot-through. The synchronous rectifiers are turned on during the freewheeling intervals, reducing conduction losses.
3.4 Snubber Circuit
In the half-bridge converter, the transformer leakage inductance can cause voltage spikes when the switches turn off. A simple RC snubber is connected across the primary winding to absorb the spike energy. The capacitance is calculated as:
$$
C_{snubber} = \frac{I_{p(max)} \cdot t_{fall}}{2 \cdot V_{peak}}
$$
The resistance is selected so that the capacitor discharges during the minimum on-time:
$$
R_{snubber} \le \frac{t_{on(min)}}{3 \cdot C_{snubber}}
$$
The power dissipation in the snubber resistor is:
$$
P_R = \frac{1}{2} \cdot C_{snubber} \cdot V_{peak}^2 \cdot f_s
$$
The actual values used are \(C_{snubber} = 1\) nF and \(R_{snubber} = 100\ \Omega\).
4. Prototype Testing and Experimental Results
4.1 Discharge Module Prototype
The discharge module prototype was tested with a power resistor as the load. Figure 1 above shows the typical operating waveforms at different discharge currents. The switching signals of the two main switches are complementary with a proper dead time. The transformer primary and secondary voltages follow the designed waveforms. Some high-frequency ringing is observed due to the transformer leakage inductance, but the snubber circuit effectively limits the voltage spikes. The battery discharge current is stable and the ripple is within the specified 5% limit.
The measured efficiency at a discharge current of 100 A is 92%, which meets the design target of 90%.
4.2 Charge Module Prototype
The charge module was tested with an electronic load and a battery simulator. At a constant charging current of 100 A, the output current waveform shows a stable average value with acceptable ripple. The transformer primary and secondary voltage waveforms are consistent with the theoretical analysis. The soft-switching performance is not fully optimized, but the overall efficiency at rated condition is 91%, which is higher than the initial requirement.
5. Conclusion
This paper has presented a comprehensive study on high-capacity EV battery pack formation equipment. The following conclusions are drawn:
- The proposed transformer series-parallel topology, based on three push-pull converters, effectively increases the output voltage level while sharing the input current evenly without extra control circuitry. The discharge module achieves over 90% efficiency at 100 A.
- The current-doubler synchronous rectification topology is well suited for the low-voltage high-current charging module. It reduces transformer winding complexity and improves efficiency, achieving 91% at 100 A.
- All system parameters, including the high-frequency transformers, filter inductors, capacitors, and snubbers, are designed using rigorous theoretical methods, and the experimental results verify the correctness of the design.
The research provides a practical solution for energy-efficient formation of large-capacity EV battery packs, contributing to both cost reduction and energy conservation in battery manufacturing.